Recently, a surface light source has come to be used more widely, not only for displays for an advertisement and personal computer, but also for liquid crystal display television sets and the like.
There is a demand of making an inverter circuit for driving these surface light sources smaller and to make high conversion efficiency.
Hereinafter, a description is given for the relation between the recent transition of an inverter circuit for a cold cathode fluorescent lamp and the invention disclosed in Japanese Patent No. 2733817 (U.S. Pat. No. 5,495,405).
For an inverter circuit for a cold cathode fluorescent lamp, conventionally, the collector resonant circuit shown in FIG. 24 has been widely used as a classical circuit. This is sometimes referred to as “Royer circuit”. However, since the Royer circuit is officially defined as a circuit which reverses a switching operation by saturating the transformer. A circuit which performs the reverse operation by using the resonance on the collector side is desirably referred to as “collector resonant circuit” or “collector resonant Royer circuit” as distinguished from the Royer circuit.
An initial inverter circuit for a cold cathode fluorescent lamp, which never uses the resonating method on the secondary side of the circuit, uses the so-called closed magnetic circuit type transformer having small leakage inductance for the step-up transformer. Under these circumstances, those skilled in the art understand that the leakage inductance of the step-up transformer in the inverter circuit is deemed disadvantageous in that it drops the output voltage on the secondary side of the transformer, and is desirable to be as small as possible.
As a result, the resonance frequency of the circuit on the secondary side of the transformer under these circumstances, without relation to the operational frequency of the inverter circuit, is set at a frequency much higher than the operational frequency of the inverter circuit in order not to influence the operational frequency of the inverter circuit. Furthermore, the ballast capacitor Cb is essential to stabilize the lamp current.
Next, among the inverter circuits for a cold cathode fluorescent lamp, the one shown in FIG. 25 is known, which is disclosed in Japanese Laid-Open Patent Publication (Kokai) No. Hei 07-211472. The circuit, where the resonance frequency of the circuit on the secondary side is three times as high as the oscillation frequency of the primary side circuit as shown in FIG. 26, has been widely used, being referred to as the so-called triple resonant circuit. In the step-up transformer used in this case, the leakage inductance is favorably made larger to some extent.
In this case, as shown in the explanatory diagram of FIG. 27, the oscillation frequency and third-order harmonic of the inverter circuit are combined to produce a trapezoidal waveform.
The actual current which flows through the cold cathode fluorescent lamp of the triple resonant circuit presents the waveform shown in FIG. 28.
The name of the step-up transformer in this case has not been fixed yet. There has been debate about whether or not it can be referred to as the “closed magnetic transformer”, which is so-called among those skilled in the art, and the definition of the name remains ambiguous. The problem of how to describe the state in which a larger amount of flux leaks although the magnetic circuit structure is closed has been discussed. There is still a problem in the lack of technical terms which consider the above state.
The shape of the transformer used in the actual so-called triple resonance is flat as shown in FIG. 29, where the flux leakage is considerably larger than in a conventional one although the magnetic circuit structure is closed. Specifically, the transformer has a large leakage inductance.
In any case, the technical idea makes the leakage inductance of the step-up transformer larger to some extent so as to form a resonant circuit between the leakage inductance and the capacitive component formed on the secondary side of the step-up transformer (FIG. 25). Also, the resonance frequency is set at a frequency three times as high as the operational frequency of the inverter circuit so as to produce the third-order harmonic in the secondary circuit (FIG. 26), thereby making the lamp current waveform trapezoidal (FIG. 27). In this case, a ballast capacitor C2, which is the ballast capacitor, operates as a part of the resonance capacitor.
As disclosed in Japanese Laid-Open Patent Publication (Kokai) No. Hei 07-211472, this technical idea considerably improves the conversion efficiency of the inverter circuit and furthermore makes the inverter circuit smaller than the step-up transformer. Also, the recent or current collector resonant inverter circuit for a cold cathode fluorescent lamp is based on the technical idea of the triple resonance, and it would not be an exaggeration to say that the technique is employed in most of the collector resonant inverter circuits which are currently used.
Next, the invention disclosed in Japanese Patent No. 2733817 (U.S. Pat. No. 5,495,405) on which the present invention is based makes the step-up transformer further smaller and improves conversion efficiency drastically. The invention, which started to be worked widely around 1996, contributes a great deal to make the inverter circuit in a laptop personal computer smaller and to improve conversion efficiency. The invention, in which the operational frequency of the inverter circuit and the resonance frequency in the secondary circuit almost coincide, is achieved by making the step-up transformer leakage inductance further larger and making the capacitive component in the secondary circuit larger at the same time in the triple resonance.
The technique utilizes an effect that the exciting current flowing through the primary winding of the step-up transformer decreases when the inverter circuit operates at a frequency close to the resonance frequency in the secondary circuit, thereby improving the power factor as seen from the primary winding side of the transformer and reducing the copper loss of the step-up transformer.
At the same time, after the invention was disclosed, as driving methods for the primary side circuit, in addition to the conventional collector resonant circuit, many kinds of driving methods including the following separately excited-type driving methods of a fixed frequency and zero current switching type driving methods for performing switching by detecting the zero current through the primary side windings. Each of these series of peripheral techniques is closely related to the invention, and contributes to popularizing usage of the resonance technique of the secondary side circuit in the invention.
Considering the changes in the series of background techniques regarding the inverter circuit for a cold cathode fluorescent lamp from a viewpoint of the leakage inductance of the step-up transformer, it can be regarded as the history that the step-up transformer leakage inductance increases and the secondary side circuit resonance frequency becomes lower at the same time as a new generation of the inverter circuit comes to the forefront as shown in FIG. 30.
It should be noted that FIG. 30 is an explanatory diagram illustrating that the relation between the drive frequency f0 of the inverter circuit and the resonance frequency fr in the secondary circuit changes with the times.
Improving the step-up transformer and selecting the drive frequency thereof appropriately achieve the miniaturization of the inverter circuit and improve conversion efficiency of the inverter circuit. Regarding this matter, in the invention disclosed in Japanese Laid-Open Patent Publication No. 2003-168585 by the inventor of the present invention (U.S. Pat. No. 6,774,580-B2), with the explanatory diagram of FIG. 31 (an explanatory diagram illustrating the improvement scheme of the power factor as seen from the driving methods side, in which the horizontal axis indicates frequency, and θ indicates the phase difference between the voltage and current of the primary winding of the step-up transformer, showing that power factor is improved as θ becomes closer to zero), a scheme for promoting the conversion efficiency as seen from the driving method side is disclosed in detail.
On the contrary, as shown in U.S. Pat. No. 6,114,814-B1 and Japanese Laid-Open Patent Publication No. Sho 59-032370, those skilled in the art consistently have advocated the technical idea that a high conversion efficiency inverter circuit is achieved by zero current switching methods.
These technical ideas, however, without having a viewpoint of the improvement power factor effect of the step-up transformer, are incorrect in that high efficiency is due to the reduction of heat generated in the switching transistor.
The reason will be described in detail below.
Zero current switching method is one power control method of the inverter circuit. A typical example thereof is a zero current switching type circuit as shown in FIG. 32, which is disclosed in U.S. Pat. No. 6,114,814-B1 and Japanese Laid-Open Patent Publication No. Sho 59-032370. The inventor of the present invention also discloses a similar technique in Japanese Laid-Open Patent Publication No. Hei 08-288080. The technique is described based on the U.S. Pat. No. 6,114,814-B1 as follows.
U.S. Pat. No. 6,114,814-B1 shows explanatory diagrams illustrating the operation of the conventional zero current switching type circuit shown in FIG. 11 which is shown as FIG. 33 in the present specification, wherein A, B show a case in which no power control is performed; C, D a case in which power control is performed; E, F a case in which zero current switching operation is tried in a state that a voltage effective value advances in phase with respect to a current value. Also, FIG. 12 of the above-mentioned U.S. Pat. No. 6,114,814-B1 is shown as FIG. 34 of the present specification, wherein G, H show one exemplary control which is not zero current switching operation.
In FIG. 33, A shows the voltage of the primary winding of the transformer when drive power is at maximum and B shows the current flowing through the transformer primary winding in that case. When the zero current switching method is used, timing when the current becomes zero is detected so as to switch driving methods. When power is at maximum, specifically when no power control is performed adjusting the duty ratio (the circulation angle) to 100%, there is no phase difference between the effective value of the effective voltage phase and the current phase supplied to the transformer primary winding. Specifically, in this condition the power factor is favorable.
Next, C of FIG. 33 shows the voltage across the transformer primary winding when the duty ratio is decreased so as to control drive power, and D shows the current flowing through the transformer primary winding in this case. In FIG. 33, the switching transistor of the driving methods is turned on at timing when the current becomes zero. On the contrary, it is not at zero current timing when the switching transistor is turned off. In this case, there is a phase difference between the effective value phase of the voltage applied to the transformer primary winding and the phase of the current flowing through the transformer primary winding. As a result, the power factor is not favorable in this case.
In FIG. 34, G shows a case in which power is controlled at a limited duty ratio in the same way so that the effective value phase of the voltage across the transformer primary winding is in phase with the phase of the current flowing through the transformer primary winding, ignoring the zero current switching method. In this case, the power factor is actually favorable as seen from the transformer primary winding side and the heat generated in the step-up transformer is small. However, this is not the result of the use of the zero current switching method.
Here, a contradiction arises in the technical idea that the zero current switching method makes higher conversion efficiency of the inverter circuit. In the technical idea of the invention disclosed in U.S. Pat. No. 6,114,814-B1, zero current switching method is eliminated in the state shown in G, H of FIG. 34, for the reason that decreasing the conversion efficiency of the inverter circuit.
It should be noted that in E, F of FIG. 34 are explanatory diagrams illustrating a case in which zero current switching operation is tried in a state that a voltage effective value advances in phase with respect to a current effective value, and G, H of FIG. 34 are explanatory diagrams showing one exemplary control which is not zero current switching operation.
According to the comparative experiments conducted by the inventor of the present invention, however, the inverter circuits have clearly higher conversion efficiency by the control method of G, H of FIG. 34 than by the control method of C, D of FIG. 33.
Consequently, the theory that the zero current switching method makes the inverter circuit higher conversion efficiency is wrong.
The background against which such a misunderstanding has occurred is as follows.
Using the zero current switching method, particularly only when no power control is performed, there is necessarily no phase difference between the effective voltage phase and the current phase of the primary winding of the step-up transformer. Therefore, the power factor of the step-up transformer is improved; the current flowing through the transformer primary winding decreases; and the current flowing through the switching transistor also decreases to a minimum. As a result, the heat generated in the step-up transformer primary winding and the heat generated in the switching transistor decrease, thereby improving the conversion efficiency of the inverter circuit. This is taken, by mistake, that the zero current switching method brings high efficiency.
In the state shown as FIGS. 11A and 11B in U.S. Pat. No. 6,114,814-B1, in which no power control is performed, the operational state thereof is equivalent to the standard current-mode resonant operational state. Specifically, as a matter of fact it is not the zero current switching method but conventional current resonant type circuit that brings the inverter circuit high efficiency.
A current-mode resonant inverter circuit is known for lighting a hot cathode fluorescent lamp, and for example, the circuit shown in FIG. 35 is generally used. In such a current-mode resonant circuit, no dimmer function is provided in its basic circuit structure. Thus, when the light output is controlled in the current-mode resonant circuit, a DC-DC converter circuit is provided at a preceding stage thereof.
FIG. 36 is an explanatory dimmer circuit of an inverter circuit for a cold cathode fluorescent lamp which combines a conventional current-mode resonant circuit, a DC-DC converter circuit at a preceding stage thereof and the leakage flux transformer invented by the present inventor (same inventor of this invention). In this example, the DC-DC converter circuit comprises a transistor Qs, an inductance Lc, a diode Ds, and capacitor Cv.
A scheme of improving the current-mode resonant circuit itself for light control has also been proposed. FIG. 37 shows the dimmer circuit disclosed by the present inventor in Japanese Laid-Open Patent Publication No. Hei 08-288080 (same inventor of this invention), in which, in a prescribed period of time after timer circuits 10, 11 detect zero current, a frequency control circuit 12 turns off switching elements 2, 3. The timer circuits 10, 11, which are RS flip-flops, are set at zero current and reset after a prescribed period of time. In this scheme, light is controlled by the method in which after the switching method is turned on by detecting zero current, the switching method is turned off.
A similar scheme is disclosed in FIG. 9 in U.S. Pat. No. 6,114,814-B1. This scheme is illustrated in the circuit diagram shown in FIG. 38 of the present specification, in which an RS flip-flop 172 is set at zero current and reset after a prescribed period of time. Both in U.S. Pat. No. 6,114,814-B1 and in Japanese Laid-Open Patent Publication No. Hei 08-288080, zero current is detected so as to turn on the switching method and to set the RS flip-flop at the same time, followed by resetting after a prescribed period of time so as to turn off the switching method. Both provide a dimmer function to the switching method in the current-mode resonant circuit, characterized in that the current delays in phase with respect to the voltage effective value when controlling light. They are based on completely the same technical ideas and their achievement methods are almost the same.
According to what the present inventor himself knows, he has confirmed that, if light is controlled based on the invention disclosed in Japanese Laid-Open Patent Publication No. Hei 08-288080, when a cold cathode fluorescent lamp or hot cathode fluorescent lamp is controlled so as to be considerably dim, a larger current flows through the transistor of the switching method thereby generating heat.
In either case, since high efficiency in the inverter circuit is clearly due to the current-mode resonant type, the present inventor has disclosed the current-mode resonant inverter circuit for a discharge lamp as FIG. 39 in Japanese Patent Application No. 2004-318059 corresponding to U.S. patent application Ser. No. 11/261,492 (same inventor of this invention).
Recently, there is a demand in making an inverter circuit high-powered in order to drive multiple cold cathode fluorescent lamps, external electrode fluorescent lamps EEFL or the like in parallel, in a liquid crystal display backlight for television for example, a number of attempts have been made to drive the inverter circuit directly using the direct-current power (generally about 400V) obtained from commercial power through a PFC circuit (power factor control circuit) as method for making the inverter circuit high-powered and reducing the cost thereof.
However, the cold cathode fluorescent lamps used for a liquid crystal display backlight for television are mostly long and their steady discharge voltages often exceed 1600V. When trying to light the cold cathode fluorescent lamps, since a conventional current-mode resonant circuit is composed of a half-bridge circuit, it is difficult to drive the inverter circuit by direct commercial power so as to light the cold cathode fluorescent lamps.
Consequently, for example, in the examples disclosed in U.S. Pat. No. 6,181,079, the cold cathode fluorescent lamps are lighted in the current-mode resonant circuit by switching the high voltage obtained after being stepped up through the step-up transformer from the PFC circuit output.
In the conventional current-mode resonant circuit, the half-bridge type has been mainly used, which is known as a lighting device for a hot cathode lamp. FIG. 40 shows one example of the inverter circuit applied in order to light a cold cathode fluorescent lamp.
However, the half-bridge circuit is less efficient in using the power supply voltage. FIG. 41 is an explanatory diagram illustrating the efficiency at which the half-bridge circuit uses the supply voltage, showing the voltage supplied to the transformer primary winding. Specifically, the reference character Er represents the waveform of the voltage at the half-bridge output stage and its effective value voltage. The reference character Es represents a rectangular (square) wave half of the power supply voltage, which would be the same if converted into the alternating voltage effective value.
Next, when power is controlled by the zero current switching method disclosed in Japanese Laid-Open Patent Publication No. Sho 59-032370 so as to control a cold cathode fluorescent lamp, the power factor is not very favorable. Furthermore, since the half-bridge configuration cannot respond to low power supply voltage, it is difficult to take full advantage of the power factor improvement effect disclosed in Japanese Patent No. 2733817 (U.S. Pat. No. 5,495,405).
As driving methods for carrying out the technical subject matter described in Japanese Patent No. 2733817 (U.S. Pat. No. 5,495,405), a separately excited-type driving method is often employed with the fixed frequency oscillation circuit composed of a capacitor C and a resistor R as an oscillation circuit. In this case, however, there are sometimes fluctuations in parasitic (stray) capacitances caused by assembly methods for mass production, thereby causing deviations with the secondary side resonance frequency circuit of the step-up transformer. Alternatively, there are sometimes fluctuations in component values thereby causing the drive frequency of the drive circuit on the primary side to deviate. In such situations, constant driving at the optimum resonance frequency at which the power factor is improved is difficult.
If the resonance frequency of the secondary side circuit is shifted away from the drive frequency of the primary side circuit, the efficiency of the inverter circuit becomes extremely worse. Therefore, when using fixed-frequency separately-excited driving methods, the Q value of the secondary side resonant circuit is lowered so as to obtain broad resonance characteristics thereby responding to frequency deviation. For such a reason, it is difficult to raise the Q value of the secondary side resonant circuit in the fixed-frequency separately-excited driving methods.
When trying to drive the secondary side resonant circuit with a low Q value by a conventional current-mode resonant circuit, continuous oscillation becomes difficult. Therefore, consideration has to be given so as not to make the Q value too low when driving by the current-mode resonant type.
However, in a general step-up transformer for a cold cathode fluorescent lamp, the Q value of the secondary side resonant circuit is never set to high. Specifically, it is because the technical idea of setting the Q value to high is not known among those skilled in the art at the time of filing of the application of the present invention.
Consequently, in order to respond to a commercial step-up transformer for fixed-frequency drive, the value of the coupling capacitor Cc on the primary side is decreased so as to resonate with the leakage inductance of the step-up transformer on the primary winding side, thereby making the coupling capacitor Cc involved in the resonance to ensure continuous oscillation with stability. However, the measures involve problems that heat is generated easily in the step-up transformer.
Next, as a method for making the inverter circuit high-powered and reducing the cost thereof, consideration is given to problems in the attempt to drive the inverter circuit directly using the direct-current power (generally about 400V) obtained from commercial power through a PFC circuit (power factor control circuit). For example, in the example disclosed in U.S. Pat. No. 6,181,079, the step-up transformer is provided after the PFC circuit so as to obtain a direct current voltage higher than 400V followed by further stepping up the direct current voltage through a half-bridge switching circuit by driving a parallel loaded serial resonance circuit so as to light a cold cathode fluorescent lamp.
However, also in this case, the half-bridge circuit, which is less efficient in using power voltage, cannot light a cold cathode fluorescent lamp directly due to its high discharge voltage.
In order that such a cold cathode fluorescent lamp can be lighted directly at DC 400V, which is the PFC circuit output, the Q value of the parallel loaded serial resonant circuit has to be set to a high value so as to make the step-up ratio higher. Specifically, the following equation shows that a large value for the Q is required in order to light 1600V cold cathode fluorescent lamps in parallel.Q=1600V(400V/2)=8At least 8 to 10 is required for the Q value in order to light the cold cathode fluorescent lamps at DC 400V.
Consequently, there has been a demand of the current-mode resonant circuit system, which uses power voltage efficiently.
Also, in the power control method by the conventional zero current switching method, power factor becomes worse when power is controlled for the following reason. In the conventional zero current switching circuit shown in FIG. 38, the relation between the voltage and current given to the primary winding of step-up transformer is exemplarily shown in FIG. 42 and FIG. 43.
The voltage waveform rises by detecting the zero point of the current. The ON timing of the switching point is at zero current, but the OFF timing thereof is not at zero current.
The voltage waveform converted into the effective value is shown with a broken line. As can be seen from FIG. 42, the current delays in phase with respect to the voltage effective value. This means that the power factor is poor. With the zero current switching method, idle current (reactive current) increases when power is controlled, thereby increasing copper loss in the step-up transformer primary winding, so that the conversion efficiency of the inverter circuit becomes worse.
Next, a description is given for the function for the power factor decreases using the zero current switching method with reference to nomographs. When using the zero current switching method, power factor is poor particularly at a narrower (smaller) duty ratio as shown in FIG. 43. This is because the current is considerably delays in phase with respect to the voltage.
A description is given in further detail as follows.
FIG. 44 shows the relation between delay angle and duty ratio, as to how considerably the current waveform delays in phase with respect to the effective voltage waveform, which is a simple inverse proportional relation.
FIG. 44 calculates how the phase of effective voltage and the current phase change along with a change in duty ratio. It is shown, for example, when the duty ratio is 25%, the delay angle of the current with respect to the voltage is 67.5 deg. From FIG. 44, the phase delay of the current with respect to the voltage when the duty ratio is set at 25% can be obtained as about 67.5 deg.
As shown in FIG. 45, in the zero current switching circuit, the intersection of the frequency corresponding to the delay angle and the phase characteristic becomes the operational frequency of the inverter circuit. In the zero current switching circuit, therefore, the operational frequency deviation is unavoidable when power is controlled.
Next, consideration is given for power factor in FIG. 46 and FIG. 47.
In FIG. 46, if the load current converted on the primary side is set to a, the exciting current is represented by tan θ, and the current through the transformer primary winding is represented by 1/cos θ (reciprocal of power factor).
FIG. 47 is an explanatory diagram showing the relation among the load current converted on the transformer primary side, the exciting current, and the current through the transformer primary winding for considering power factor. FIG. 47 illustrates that a large delay angle allows a larger exciting current thereby increasing idle current.
In FIG. 47, the combined current ratio represents 1/cos θ (reciprocal of power factor). Taking the current delay in phase with respect to the voltage effective value as a current delay angle θ, the figure shows its relation with 1/cos θ (reciprocal of power factor). How much large current flows through the transformer primary winding than load current is considered in FIG. 47 as follows. If the current delays by 67.5 deg. in phase with respect to the voltage effective value, the current flows through the transformer primary winding is 2.61-times larger than in a case in which there is no delay. Consequently, the power factor becomes extremely worse, and more heat is generated in the transformer primary winding due to increase in copper loss. Also, for the same reason, more heat is generated in the transistor of the switching method.
Specifically, when power is controlled using the zero current switching method, if using the duty ratio control method disclosed in each of U.S. Pat. No. 6,114,814-B1, Japanese Laid-Open Patent Publication No. Hei 08-288080 and Japanese Laid-Open Patent Publication No. Sho 59-032370 for power control, the following conclusion is obtained from a viewpoint of improving power factor.
In a state that the duty ratio is large, specifically, in a state that the current slightly delays in phase with respect to the voltage effective value, the conversion efficiency of the inverter circuit is favorable. However, when the duty ratio is small, the current delays considerably in phase and consequently, the power factor becomes worse, and a larger current flowing through the transformer primary winding makes the inverter circuit conversion efficiency worse. Particularly, as the duty ratio becomes smaller thereby delaying the current in phase closer to 90 deg., idle current increases rapidly thereby making the efficiency significantly worse.
Specifically, in such a state, when the zero current switching method is applied to a laptop personal computer, if an AC adapter, the supply voltage becomes the largest. Under these conditions, when power is restricted so as to make a liquid crystal display panel darker or the like, the current delays longest in phase. In this case, significant heat is generated in the inverter circuit in practice.
Furthermore, there is also a problem that the operational frequency deviation of the inverter circuit is unavoidable when current is controlled by the zero current switching method.
What is clear is that the technical idea of the zero current switching is not always necessary in order to configure the high-efficiency inverter circuit in a state that power is controlled. On the contrary, the idea is damaging. In order to configure an inverter circuit with good conversion efficiency, the above technical idea has to be eliminated and a method in which the power factor becomes best in the step-up transformer primary winding has to be applied.